A conventional spectrum spread receiver will be described here. Conventional spectrum spread receivers that use the CDMA system which adopts the spectrum spread modulation system have been disclosed in, for example, “Experimental Evaluation on Coherent Adaptive Array Antenna Diversity for DS-CDMA Reverse Link, The Institute of Electronics, Information and Communication Engineers, Technical Report of IEICE, RCS98-94 p.33-38, September, 1998”, and “Laboratory Experiments on Coherent Rake Receiver in Broadband DS-CDMA Mobile Radio, The Institute of Electronics, Information and Communication Engineers, Technical Report of IEICE, RCS99-129 p.57-62, October, 1999”.
A construction and an operation of the conventional spectrum spread receiver(s) disclosed in the above-mentioned references will explained now. FIG. 6 shows the construction of the conventional spectrum spread receiver. In FIG. 6, reference numerals 500, 501, . . . , 502 are antennas the number of which is represented by N (a natural number), 510, 511, . . . , 512 are band-path filters (BPFs), 520, 521, . . . , 522 are reverse spread sections, 530, 531, . . . , 532 are beam forming sections for individually forming beams from L (a natural number) paths generated based upon a signal after the reverse spread that has been subjected to an influence from a multi-path waveform, 540 is a path detector, 550, 551, . . . , 552 are complex multipliers, 553 is a delay unit, 560 is a weight controlling section, 561 is an adder, 562 is a complex multiplier, 563 is a complex conjugate calculator, 564 is a subtracter, 565 is a complex multiplier, 570 is a transfer path estimating section for estimating the transfer path with respect to each of individual paths, 580, 581, . . . , 582, . . . , 583 and 584 are delay units, 585 is an adder, and 590 is a data determining section.
An operation of the conventional spectrum spread receiver will now be explained. First, signals received from a mobile station by the N antennas 500, 501, . . . , 502 are filtered by the BPFs 510, 511, . . . , 512, and subjected to desired band-width limitations. The signals after having been subjected to the band-width limitations are input into the reverse spread sections 520, 521, . . . , 522 at which they are subjected to reverse spreads by the same sequence as the spread code sequence (corresponding to the PN sequence) that have been used on the transmission side.
The path detector 540 selects L paths from a specific one of the signals that have been subjected to the reverse spread signal that has been influenced by the multi-path wave. A detailed explanation will be given of the operation of the path detector 540. FIG. 7 shows the construction of the path detector 540. In FIG. 7, reference numeral 600 is a transfer path estimating section, 601 is an average power-value calculator, 602 is a threshold value calculator, 603 is a judgment section and 604 is a path selector.
In this path detector 540, first, the transfer path estimating section 600 adds all the symbols within one slot in the same phase based upon a pilot symbol (known signal) placed at a slot unit, and outputs a spontaneous transfer-path estimating value as a result. Subsequently, the average power value calculator 601 carries out an averaging operation of power over several slots by using the received transfer path estimating value, thereby calculating the average power delay profile as the result of the operation.
In the threshold value calculator 602, among the received average power delay profiles, the path having the smallest power is regarded as noise or interference power, and the power value that is greater than the power by ΔdB is output as a threshold value used for the path selection. Then, the judgment section 603 compares the average power delay profile and the threshold value, and all the paths having average power values greater than the threshold value are set as the multi-paths corresponding to desired signals. Further, it outputs the time-sequential positional information of these paths and the power values of these paths.
In the path selector 604, since each beam forming section carries out a signal processing only on L paths preliminarily determined due to the limitations of H/W and S/W, the L paths are selected in the descending order from the greatest average power value. Thus, the time-sequential position corresponding to each path is output as the path position information. FIG. 8 shows the processes in the threshold value calculator 602, the judgment section 603 and the path selector 604.
After the output of the path positional information from the path detector 540, the beam forming sections 530, 531, . . . , 532 form beams by signal processes based upon an applicable algorithm. The beam forming section 530 is used for carrying out signal processing on the path having the greatest signal power, and the beam forming sections 531, . . . , 532 are used for carrying out signal processing on the paths having the second greatest signal power to the L-th greatest signal power. The following description will discuss the operation of the beam forming section 530 in detail.
As described above, the reverse spread signal from the reverse spread section 520 is separated by the path detector 540 into each path unit, and input into the beam forming section 530. Therefore, in each beam forming section, the beam is formed on a path unit basis that has been detected.
First, in the weight controlling section 560, the calculation of weight is carried out based upon an adaptive algorithm such as LMS (Least Mean Square), and in each of the complex multipliers 550, 551, . . . , 552, the signal received by each antenna is multiplied by a complex weight for forming a beam on the basis of a path. Then, the adder 561 combines the respective receiving signals that have been multiplied by the complex weights, and outputs the results of the combined as an antenna combined signal having directivity.
Next, the transfer path estimating section 570 estimates the transfer path. More specifically, for example, by using pilot symbols of a known sequence that are provided for the respective slots, a transfer path estimation value (complex value) with respect to the first path is calculated. FIG. 9 shows the slot construction.
The complex conjugate calculator 563 calculates the complex conjugate value of the transfer path estimation value calculated in the transfer path estimating section 570. Then, the complex conjugate value is input into the complex multiplier 562 in which it is multiplied by the antenna combined signal, thus, a weighting process is carried out in proportion to the signal amplitude, and a signal from which a phase variation has been removed is output.
After the beam forming sections 530, 531, . . . 532 have formed the first (the path having the greatest signal power) to the L-th beams (the path having the L-th greatest signal power), the delay units 580, 581, . . . , 582 respectively add amounts of delay D1, D2, . . . , DL thereto so that all the paths from the first path to the L-th path have the same timing.
The adder 585 adds the signals which have been allowed to have the same phase on the basis of a path. The data decision section 590 performs a hard determination on the data. The result of the hard determination is output as demodulation data of the receiver. Here, since the results of the hard determination are used as reference signals for forming the beams of the respective paths, the delay units 583 to 584 respectively carry out delay adjustments so that, for example, amounts of delay, DL−D1, DL−D2, . . . , 0 (where the L-th path is not subjected to a delay) are added thereto.
An explanation will be given on how the weights to be added to the respective receiving signals is determined by, for example, the beam forming section 530. It will be assumed here that, an already known algorithm is used for forming the beams.
For example, the output of the delay unit 584 is multiplied by the transfer path estimation value in the complex multiplier 565, to form a reference signal. Thereafter, in the subtracter 564, the antenna combined signal is subtracted from the reference signal to generate an error signal e1 (k) to be adaptive to the first path. Then, the weight controlling section 560 updates/determines the weight in accordance with equation (1) that indicates the normalization LMS.                                           W            1                    ⁡                      (                          k              +              1                        )                          =                                            W              1                        ⁡                          (              k              )                                +                      μ            ⁢                                                   ⁢                                                            X                  1                                ⁡                                  (                                      k                    -                    τ                                    )                                                                                                                                          X                      1                                        ⁡                                          (                                              k                        -                        τ                                            )                                                                                        2                                      ⁢                                          e                1                            ⁡                              (                k                )                                                                        (        1        )            
Here, the denominator of the second term on the right side of equation (1) represents a norm, k represents the sampling time (t=kTS: TS is a sampling cycle), X1(k) is a vector expression (X1(k)=[x1(1, k), x1(2, k), . . . . , x1(N, k)]T) of the first path of each reverse spread signal, and W1(k) is a vector expression of each weight with respect to the first path (w1(1, k)=w1(1, k), w1(2, k), . . . , w1(N, k)T). Moreover, the initial value w1(0) of W1(k)=[1, 0, . . . , 0]T, μ represents the step size, and τ represents a delay time (amount of delay).
In this manner, in the conventional spectrum spread receiver, with respect to L paths detected from the receiving signals received from a plurality of antennas, beams are individually formed (by using adaptive algorithms), that is, the SIR (signal to interference power ratio) with respect to a desired signal is improved while directing a null set to the interference signal by carrying out a weighting combining (Rake combining) in accordance with the transfer path estimation value. Moreover, the conventional spectrum spread receiver is allowed to have an optimal channel capacity in the case when the positional distribution of mobile stations within a cell to which a base station can provide services is uniform and in the case when the beam interference power of beams formed on the basis of a path is the same.
However, in the above-mentioned conventional spectrum spread receiver, in the case when the interference wave power of beams formed on the basis of a path is not regarded as the same due to the fact that the positions of mobile stations are instantaneously biased or the fact that mobile stations having different transmission signal powers exist because of different transmission speeds, the SIR is not optimized, failing to obtain a superior bit error characteristic; consequently, the resulting problem is that it is not possible to obtain an optimal channel capacity.
Moreover, in the conventional spectrum spread receiver, another problem is that, in the case when a mobile station that is a subject for communication is shifted and the shifting velocity is a high speed, it is difficult for the base station to direct beams to the mobile station with high precision.
Moreover, in the initial state for forming beams by using an adaptive array antenna in the conventional spectrum spread receiver, since it is difficult to tell the arrival direction of multi-path waves from a mobile station to the base station, and since it is not possible to form a beam having a sharp directivity, the selection of the path is carried out by utilizing a single antenna as described above. However, the resulting problem is that, in the case of using the single antenna, it is not possible to detect paths with high precision, in a transfer path under a great effect of interference.
Moreover, in the conventional spectrum spread receiver, as described above, in the case when a single antenna is used, the weight is set to each receiving signal. In this case, however, a long period of time is required until the beam has been formed based upon the adaptive algorithm, and on the transmission side of the mobile station, much transmission signal power is required until the beam formation has been finished, in order to satisfy the predetermined quality on the base station. Consequently, an instantaneous increase occurs in the interference power, causing a failure to obtain an optimal channel capacity.
It is an object of the present invention to provide a spectrum spread receiver which can achieve a desirable bit error rate characteristic, even in the case when the interference powers of the beams formed on the basis of a path are not considered to be the same.
It is an another object of the present invention to provide a spectrum spread receiver which, even in the case when a mobile station that is an object of communication is shifted, and when the shifting velocity is high, allows the base station to direct a beam to the mobile station with high precision.
It is a still another object of the present invention to provide a spectrum spread receiver which, even in the case when a path is selected by using a single antenna, can carry out the path detection with high precision, and also can greatly shorten the time required for forming beams based upon an adaptive algorithm.